Dual-polarized Dual-band Mobile 5G Antenna Array

Igor Syrytsin, Shuai Zhang and Gert F. Pedersen

Department of Electronic Systems, Aalborg University, Fredrik Bajers Vej 7, Aalborg, Denmark

Keywords:

Antenna Array, 5G Antenna, Dual-band, Dual-polarized Antenna, Phased Array.

Abstract:

In this paper, a dual-band dual-polarized phased antenna array for 5G mobile terminals is proposed. The array

has a bandwidth of 3.6 GHz and two resonances at 30.5 and 32.8 GHz. The array has a clearance of 2.85 mm

and fed with two ports in simulation to excite the notch and dipole parts of the proposed antenna structure.

Finally, the proposed antenna element is combined into two arrays with different conﬁgurations. It is shown

that it is better to use the array of 8 elements than to use two 4-element sub-arrays with orthogonal orientation.

1 INTRODUCTION

In the recent years the research community and the

industry has been working towards standardization

of 5G mm-wave communication system. Because

the bandwidth is a scarce resource at the current fre-

quency bands under 6 GHz it is decided to implement

the 5G mm-wave system in the mm-wave frequency

spectrum (Rappaport et al. 2013). Currently, eleven

candidate bands in the range between 24.25 GHz and

86 GHz have been considered for the 5G mm-wave

communication system (Lee et al. 2018). To combat

the high path loss expected at the mm-wave frequen-

cies antennas with the gain higher than 7 dBi at both

mobile and base stations. However, because the ori-

entation of the mobile terminal is not known, beam-

forming will be implemented in order to achieve spa-

tial coverage requirements (Roh et al. 2014). The spa-

tial performance of the mobile terminal can be char-

acterized by using the metric of coverage efﬁciency,

which has been ﬁrst proposed in (Rehman et al. 2012)

and then applied to 5G mobile terminal antennas in

(Helander et al. 2016). Furthermore, in (Nielsen &

Pedersen 2016) it has been shown that in the typ-

ical indoor propagation channel the received power

depends strongly on the polarization of transmitter

and receiver antennas. The measurements has shown

that up to 10 dB difference can be seen between dif-

ferent antenna polarization combinations, but the co-

polarized antenna conﬁgurations are not always the

best option for the indoor channel. Thus, a polariza-

tion reconﬁgurable antenna is required to adapt to the

channel changes and keep the received power at the

highest possible level.

Antenna array with multiple polarizations is pro-

posed in (Hong et al. 2015) for the mm-Wave 5G

mobile terminals. Then, a low-proﬁle antenna solu-

tion with beam-steering capabilities is proposed in in

(Hong et al. 2014). A Vivaldi phased antenna array

performance and it’s user effects are investigated in

(Ojaroudiparchin et al. 2015). In (Hussain et al. 2017)

a compact 4G MIMO antenna is integrated with the

5G mm-wave mobile array. Two different methods in

(Ojaroudiparchin et al. 2016) and (Zhang et al. 2017)

are introduced in order to create a 3D coverage 5G

mm-wave phased antenna array system. Three sub-

arrays mounted on the folded 3D structure are con-

structed in (Ojaroudiparchin et al. 2016). Then, in

(Zhang et al. 2017) a surface wave is efﬁciently uti-

lized to change the radiation direction of slot array

elements. However, for 5G mm-wave a bandwidth of

at least 1.6 GHz is required. The wideband antenna

array for 5G mobile terminals has been presented in

(Syrytsin et al. 2018). The proposed antenna element

utilizes four modes in order to achieve the wideband

performance. Circular polarized antennas has been

proposed in (Mahmoud & Montaser 2018), (Syrytsin

et al. 2017) and (Shuai Zhang 2018). Furthermore,

polarization reconﬁgurability and small clearance are

also very important design considerations for 5G mo-

bile antennas.

In this work, a dual-band dual-polarized 5G mo-

bile phased antenna array is presented. The proposed

antenna consists of two co-located antennas which

can operate at the same frequency but radiate with

orthogonal polarizations. The antenna structure is de-

signed for the 5G frequency band of 30.8 to 33.4 GHz.

However, because the 5G frequency bands are not ﬁ-

nally deﬁned yet, it has been chosen to increase the

bandwidth of the antenna by introducing a second res-

Syrytsin, I., Zhang, S. and Pedersen, G.

Dual-polarized Dual-band Mobile 5G Antenna Array.

DOI: 10.5220/0006919403090315

In Proceedings of the 15th International Joint Conference on e-Business and Telecommunications (ICETE 2018) - Volume 1: DCNET, ICE-B, OPTICS, SIGMAP and WINSYS, pages 309-315

ISBN: 978-989-758-319-3

309

onance. A bandwidth of 3.6 GHz is achieved by both

antennas with the ground plane clearance of 2.85 mm.

The antenna structure can be easily tuned to other

frequency range by changing a number of antenna

structure dimensions. Finally, the performance of the

phased array in two conﬁgurations has been investi-

gated. The metrics of the total scan pattern and cov-

erage efﬁciency have been used to quantify the simu-

lation results.

2 ANTENNA ELEMENT

PERFORMANCE

In this section, the geometry, operation principle, per-

formance and design considerations of the proposed

antenna element will be described. Surface currents,

reﬂection coefﬁcient and radiation patterns are used

to describe the performance of the antenna element.

2.1 Antenna Geometry

The geometry of the proposed antenna element is

shown in Figure 1. The antenna is built on the

Rogeres RO4350B substrate with a thickness of

0.762 mm. As shown in Figure 1, the dual polarized

antenna element consists of two parts. In simulation

setup, the dipole part is fed by the port P1, which is lo-

cated between the two dipole arms. The other port is

located between the top and bottom layer of the PCB.

The port 2 (P2) induces the currents on the top and

bottom rings around the patch in the middle, and thus

produce the radiation. In Figure 1 the constant geom-

etry dimensions are shown as numbers, and variable

dimensions are displayed in words. The variable di-

mensions can be altered in order to change the reso-

nant frequency of the antenna modes.

2.2 Antenna Operation Principle

The reﬂection coefﬁcients at the ports 1 and 2 are

shown in Figure 2 and denoted as s-parameters S11

and S22. It can be noticed that two resonances ap-

pear when the antenna structure is fed at either port 1

or port 2. A resonance frequency of the ﬁrst mode is

around 30.5 GHz and around 32.8 GHz for the second

mode. Furthermore, in Figure 2 a band from 31.8 to

33.5 GHz is visualized in a gray color.

To grasp the operation principle of the proposed

antenna it has been chosen to show the maximum sur-

face currents on the antenna structure in Figure 3. The

substrate and the bottom layer are hidden. Notice the

orientation of Z-axes in the ﬁgure, which is pointing

outside of the paper. From both ﬁgures it is clear that

(a)

(b)

Figure 1: Geometry of the proposed antenna element (a)

front view, (b) back view.

Figure 2: Reﬂection coefﬁcients at the port 1 and port 2.

the vias can efﬁciently reduce the surface wave on

the ground plane. The dipole is excited by the port

1 in Figure 3(a) at 30.5 GHz and in Figure 3(a) at

32.8 GHz. It can clearly be seen that in mode 1 the

radiation is created by the left dipole arm, and mode 2

is created by the currents running on the right dipole

arm and stub. In comparison to the dipole modes,

the modes of the rings induce higher currents on the

ground plane in Figure 3(c) and Figure 3(d). When

port 2 is excited at 30.5 GHz then the highest cur-

rents are concentrated on the ring (left and right side).

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310

(a) (b)

Figure 3: Maximum surface currents on the antenna struc-

ture produced by the (a) port 1 – mode 1, (b) port 1 – mode

2, (c) port 2 – mode 1, and (d) port 2 – mode 2.

(a) (b)

Figure 4: Radiation pattern of the proposed antenna struc-

ture excited at (a) port 1 - yz-polarization, (b) port 1 - xz-

polarization, (c) port 2 - yz-polarization, and (d) port 2 -

xz-polarization.

However, when port 2 is excited at 32.8 GHz, then the

currents are equally distributed among the ring and

ground plane. Furthermore, the patch in the middle,

between dipole arms, is added to tune the impedance

matching of a dipole. Thus, the patch is not excited

when the ring is radiating instead of the dipole.

Figure 5: Realized gain over the frequency range of the pro-

posed antenna.

2.3 Antenna Performance

Next, it has been chosen to show radiation patterns

of the antenna structure excited with port 1 and port

2. Here, two distinct polarizations are deﬁned: xz-

polarization and yz-polarization. The radiation pat-

tern of the proposed antenna structure is shown in

Figure 4. It can clearly be seen that the dipole antenna

structure has yz-polarization, as shown in Figure 4(a).

When the structure is excited by the port 1 the xz com-

ponent of the antenna gain is very low in Figure 4(b).

However, for when the structure is excited by the port

2 the xz component of the antenna gain has a highest

value in Figure 4(d).

Next, the maximum gain over the frequency range

from 24 to 40 GHz is shown in Figure 5. The realized

gain in the band of interest is higher than 2 dBi when

the antenna structure is excited by either port 1 or 2.

2.4 Dipole Design

In this section, it will be shown how to control the

resonance behavior of the dipole part of the proposed

antenna structure (when the structure is excited by the

port 1). The parametric results for the reﬂection co-

efﬁcient at port1 are shown in Figure 6 for the three

different antenna structure parameters deﬁned in Fig-

ure 1. First, the length of one of the dipole arms l

dip

changed from 0.4 to 1.1 mm as shown in Figure 6(a).

It can be seen that by changing that parameter the

matching of both modes is changing, but the reso-

nance frequency of modes remains the same. Next, in

Figure 6(b) the length of another dipole arm is swept

from 0.1 to 1.1 mm. Here more severe effect on the

resonance 1 is observed, but resonance frequency of

mode 2 remains the same. Finally, the length of the

stub is swept from 0.5 to 1 mm. Now the resonance

frequency of mode 2 changes signiﬁcantly, while the

resonance frequency of the mode 1 is unchanged. To

move the antenna resonances one should ﬁrst change

the resonance of mode 1 by changing the length l

dip2

Dual-polarized Dual-band Mobile 5G Antenna Array

311

(a)

(b)

(c)

Figure 6: Parametric results of the reﬂection coefﬁcient

when lengths (a) l

dip

, (b) l

dip2

, and (c) l

stub

are permuted.

, then change the resonance of mode 2 by tuning the

length of stub l

stub

, and ﬁnally match the antenna ac-

cording to the speciﬁcations by tuning the parameter

dip

2.5 Notch Design

In this subsection it will be shown how to control

the resonance behavior of the notch part of the an-

tenna structure (when the antenna structure is excited

by the port 2). The corresponding antenna structure

parameters are deﬁned in Figure 1. To tune the res-

onance frequencies of notch modes it has been cho-

sen to show the effect of changing the values of four

parameters of the antenna structure which is shown

in Figure 7. First, it can be noticed that a single pa-

rameter cannot be used to change the matching of the

antenna. The resonance frequency of the modes al-

ways shifts, so multiple notch parameters need to be

adjusted in order to change the matching of the an-

tenna. To change the resonance frequency of mode

2 the parameter l

notch

and w

notch

should be permuted

in Figure 7(a) and Figure 7(b). However, if the reso-

nance frequency of mode 1 is to be altered, then the

parameters w

ring

, w

cut

, and w

notch

should be permuted.

It can be already noticed that the design of the notch

part of the antenna is more complicated and not so

straightforward as the dipole design.

3 PERFORMANCE OF THE

PHASE ARRAY

In this section, the performance of the mm-wave

phased array constructed by using the proposed dual-

polarized antenna element will be investigated. To in-

vestigate the performance of the array the metrics of

the total scan pattern and coverage efﬁciency are used.

The coverage efﬁciency is calculated from the total

scan pattern (TSP) of the phased array or switchable

antenna array system and obtained from all antenna

array patterns, corresponding to the different scan an-

gles. The best achievable gain is extracted at every

spatial point.

The coverage efﬁciency is deﬁned as (Helander

et al. (2016)):

Coverage Solid Angle

Maximum Solid Angle

(1)

where the maximum solid angle deﬁned as 4π stera-

dians. The coverage efﬁciency has no unit and varies

from 0 to 1 (corresponding to 0 and 100 % coverage).

3.1 Array Geometry

In this paper, it has been chosen to investigate the

performance of the proposed antenna element in the

linear arrays in two conﬁgurations. The two array

conﬁgurations are shown in Figure 8. In the con-

ﬁguration 1 in Figure 8(a) 8 elements are distributed

into two sub-arrays of four elements, perpendicular to

each other. And in the conﬁguration 2 in Figure 8(b)

8 same elements are combined into one linear array of

8 elements. The point is to investigate which conﬁgu-

ration has better performance. From the system point

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312

(a)

(b)

(c)

(d)

Figure 7: Parametric results of the reﬂection coefﬁcient

when lengths (a) l

notch

, (b) w

notch

, (c) w

ring

, and (d) w

cut

are permuted.

(a)

(b)

Figure 8: Two conﬁgurations of the phased antenna array

(a) two sub-arrays of four elements and (b) one 8-element

phased array.

(a) (b)

Figure 9: Total scan patterns of (a) conﬁguration 1 – xz

polarization, (b) conﬁguration 1 – yz polarization, (c) con-

ﬁguration 2 – xz polarization, and (d) conﬁguration 2 – yz

polarization.

of view, the conﬁguration 1 requires less complicated

feeding network and SPDT switch. Where the conﬁg-

uration 2 requires more complicated feeding network

but no switch. Furthermore, the sub-arrays in the con-

ﬁguration 1 is oriented perpendicular to each other in

order to maximize the coverage (scanning in xz and

yz-planes). On the other hand, the phased array in the

conﬁguration 2 can only scan in yz-plane.

The total scan patterns of the two proposed phased

antenna array conﬁgurations at 32 GHz are shown in

Fig. 9. It can be seen that the maximum gain in the

conﬁguration 1 in lower than the maximum gain in

the conﬁguration 2. However, side lobes are higher in

the case of conﬁguration 2 in Figure 9(d).

Dual-polarized Dual-band Mobile 5G Antenna Array

313

(a)

(b)

Figure 10: Coverage efﬁciency of phased antenna array in

(a) conﬁguration 1 and (b) conﬁguration 2.

Finally, the coverage efﬁciency of the array in two

conﬁgurations is calculated and shown in Figure 10.

First, it can be noticed that difference in the coverage

between two polarization is very small for the gains

lower than 0 dBi. However, a bigger difference can

be seen as the gain increases. Finally, the curves for

the array conﬁguration 1 and conﬁguration 2 have a

very similar slope, but the absolute level of the cov-

erage is different. It can be seen that the array in the

conﬁguration 1 have 20 % less coverage for the gain

of 5 dBi.

4 CONCLUSION

In this paper, a dual-polarized dual-band phased an-

tenna array for 5G mobile devices has been presented.

The proposed array have the bandwidth of 3.6 GHz

and covers the band of 30.8 to 33.4 GHz. Further-

more, it has been shown how to tune each mode of the

antenna in order to achieve the desired bandwidth and

resonance frequency of each mode. Next, two phased

array conﬁgurations have been investigated. One con-

ﬁguration has 8 array elements distributed into two

4-element sub-arrays oriented orthogonally with re-

spect to each other. It has been found that array of

8 elements give better spatial coverage performance.

However, this investigation has only been done for the

broadside antenna element. A further investigation of

endﬁre antenna element should also be conducted in

the future work.

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